In recent small-sized domestic products and office automation (OA) equipment, high-frequency inverters have been mounted for achieving high performance and high efficiency.
Also, in fluorescent lamp appliances for homes and fluorescent lamp appliances for facilities, a copper-iron type ballast, such as a choke current-limiting type, a leakage transformer type or the like, has hitherto been used as a circuit system that drives a fluorescent lamp. However, since it has limitations on the aspects of shape, weight, and efficiency, a lamp controller called a high frequency lighting type ballast (inverter type ballast) comes into use in the present fluorescent appliances, and is also being used in HID lamp (mercury lamp, metal halide lamp, etc.) appliances, bulb type fluorescent lamps, etc.
This inverter type ballast has advantages in that it has high efficiency and is able to save electric power, and it is also able to reduce lamp flickering and ballast noise, and furthermore, it is able to reduce its weight. For these reasons, inverter control of the above-mentioned fluorescent lamp appliances has made rapid progress.
However, in the above-mentioned high-frequency inverter or inverter type ballast (hereinafter referred to as an "inverter"), a capacitor smoothing circuit method for full-wave rectification, which employs rectifiers (diodes) and performs smoothing with an electrolytic capacitor, is often used, and distorted-wave current resulting from diode nonlinearity flows in a commercial power supply.
For that reason, a harmonic component (harmonic current) flows in an input current on the side of a commercial power supply. The problem of failure (harmonic failure) that is affected by this harmonic current has become significant.
For this reason, circuit techniques for suppressing harmonic current have been studied and investigated. For instance, an AC reactor insertion method, a partial smoothing method, an active smoothing filter method (see "Inverter Fluorescent Lamp," Electronic Technique, Vol.32, No.3, pp.113-119), a dither rectifying method (see "High Power-Factor Switching Regulator Employing Dither Effect," National Convention Lecture Thesis Collection of Electric Society, No. 546, pp. 5-137), etc., have been proposed.
Furthermore, as an electronic ballast for fluorescent lamps, a neutral-point electronic ballast circuit has been proposed in which a reduction in the harmonic component of an input current on the side of a commercial power supply is performed with only an inverter for lighting a fluorescent lamp, as in the dither rectifying method (see Yoshihito Kato, "One Method of a Simple Harmonic Reduction Circuit," Electrical Equipment Society Journal, Vol. 12, No. 10, pp. 902-904). A study of the theoretical analysis of this neutral-point electronic ballast circuit (neutral point inverter type ballast) has also been made (see Yoshihito Kato, "Development of an Input Current Low-Distortion Type Electronic Ballast by a Neutral Point Inverter," Illumination Society Journal, Vol. 79, No. 2, pp. 14-20).
This neutral point inverter type ballast has many advantages in that (1) by inserting a low-pass filter LPF in the side of a commercial power supply, a reduction in the harmonic component contained in an input current is possible with only an inverter for lighting a fluorescent lamp, as in the active smoothing filter method, (2) there is no need to make a new circuit as in the dither rectifying method, and this ballast is applicable to an improvement in the existing half-bridge ballast, (3) the harmonic component of an input current can be reduced to less than IEC standard (IEC 1000-3-2), (4) for an input power factor, a high power factor of 97% or more is obtained, (5) circuit constitution is simple and also a reduction in the luminous efficiency of the lamp is low, and soon. For these reasons, the neutral point inverter type ballast is being used as a suitable circuit that prevents the harmonic failure of an inverter.
FIG. 19 is a basic circuit diagram of a neutral point inverter. This circuit consists of a full-wave rectifier DB that rectifies a commercial power supply Vi to direct-current voltage Ed through a low-pass filter LPF (the constituent diodes in the circuit diagram are represented simply as 1 through 4, and in the specification, they are referred to as DB1 through DB4.), a smoothing capacitor Cs that smooths the output of the full-wave rectifier DB, a series circuit that is connected in parallel with the smoothing capacitor Cs and also consists of voltage-dividing capacitors C1 and C2 for dividing direct-current voltage Ed, a series circuit consisting of switching elements Q1 and Q2 connected in parallel with the smoothing capacitor Cs, and load RL connected between the connecting point which is between the voltage-dividing capacitors C1 and C2 (hereinafter referred to as a "neutral point") and the connecting point which is between the switching elements Q1 and Q2 (hereinafter referred to as a "SW point"). The neutral point is connected to one end of a commercial power supply Vi.
For the operation of this circuit, the ripple voltage included in the output of the full-wave rectifier DB is converted to direct-current voltage Ed with the smoothing capacitor Cs. Then, the switching elements Q1 and Q2 are turned on or off to constitute a closed circuit that includes the neutral point. With the closed circuit, the voltage-dividing capacitor C1 or C2 is charged from the smoothing capacitor Cs. This charging current becomes load current that flows in the load RL, and reverse current is ensured for an interval during which no load current flows. If the switching elements Q1 and Q2 are alternately turned on and off (inverting operation), voltage VL with a high frequency superposed on a commercial frequency will be applied across the load RL. Since the current through the diodes DB1 through DB4 has a triangular high frequency with a quiescent interval proportional to load, the current is passed through the low-pass filter LPF to obtain a false sine current waveform. This makes a reduction in the harmonic component of the input current on the commercial power supply possible.
FIG. 20 is a circuit diagram of the case where a fluorescent lamp LT is employed as the load of a neutral point inverter, this circuit being called a neutral point inverter type ballast. Since the load voltage VL that is obtained with only the basic circuit (FIG. 19) is a charging-discharging (particularly charging) waveform from both the switching elements Q1 and Q2 and the voltage-dividing capacitors C1 and C2, the load voltage VL is unsuitable for lighting of the fluorescent lamp LT. In order to remove this transient portion and in order to make the lamp current a sine wave, a series circuit, which consists of an inductor L1 and a fluorescent lamp LT, is connected to the load terminal of the basic circuit (between the neutral point and the SW point), and a resonance capacitor is connected in parallel with the fluorescent lamp LT so that it resonates with the fluorescent lamp LT. This circuit constitution (load circuit) is the circuit shown in FIG. 20 (hereinafter referred to as an "implementation circuit"). A description will hereinafter be given of the operation of this implementation circuit. When a smoothing capacitor Cs has a sufficiently larger value than voltage-dividing capacitors C1 and C2 (Cs&gt;&gt;C1, C2), the voltage of the maximum value Vm of an input voltage (Vi=Vm.multidot.sin(wt)) is obtained across the smoothing capacitor Cs (Vm=Ed). This is because although the rectifier DB and the voltage-dividing capacitors C1 and C2 apparently constitute a voltage doubler circuit, the influence of C1 and C2 becomes negligible if Cs is sufficiently larger than C1 and C2 and therefore a voltage doubler circuit is not constituted.
Therefore, in a steady state which performs an inverting operation, if it is assumed that voltages across the voltage-dividing capacitor C1 and across the capacitor C2 are respectively V.sub.C1 and V.sub.C2, the voltage Ed across the smoothing capacitor Cs is expressed as follows: EQU Ed=V.sub.C1 and V.sub.C2
On the other hand, in this implementation circuit, with the on-off operation of the switching elements Q1 and Q2, charging-discharging current flows in both the voltage-dividing capacitors C1 and C2 and the smoothing capacitor Cs. Also, when an input voltage Vi is Ed&lt;.vertline.Vi.vertline., charging current from a commercial power supply is superposed on the smoothing capacitor Cs, and the input current Ii has approximately a so-called capacitor-input current waveform. Therefore, it is considered that the input current Ii has a pointed waveform.
With this, the waveforms of the voltages V.sub.C1 and V.sub.C2 across the voltage-dividing capacitors C1 and C2 become waveforms such as those shown in FIGS. 21 and 22, respectively. Also, the voltage VR that is generated in a load circuit R becomes a composite waveform such as that shown in FIG. 23, in which FIGS. 21 and 22 showing the voltage waveforms across the voltage-dividing capacitors C1 and C2 are superposed with alternating current zero as a boundary. Note that since FIG. 23 has been simplified, the actual voltage VR is one in which high-frequency voltages from the maximum of V.sub.C1 to the minimum of V.sub.C2 are applied to the load circuit R.
Next, examine the input current Ii. As evident in the above-mentioned description, in the time period while Ed&lt;.vertline.Vi.vertline., a current for charging the smoothing capacitor Cs flows in the smoothing capacitor Cs. This charging current goes to a greater value, unlike current flowing in the voltage-dividing capacitors C1 and C2, depending upon the states of the switching elements Q1 and Q2 in the time period while 0&lt;.vertline.Vi.vertline.&lt;Ed.
With this, if the waveform of the input current Ii is shown in a figure, it becomes an intermittent current interrupted by the switching elements Q1 and Q2. The input current Ii also has a current waveform which becomes discontinuous at a position in which the input voltage Vi crosses zero, the current waveform having a peak such as that shown in FIG. 24. Therefore, if the low-pass filter LPF for passing a commercial frequency is inserted in an input, a current waveform which is nearly a sine wave will be obtained and therefore high-frequency current will be prevented from flowing in a commercial power supply. However, slight wave distortion is produced by the peak current. For this reason, there is a need to select an optimal smoothing capacitor Cs so that this peak current is reduced.
The above-mentioned description has been made with reference to the case where the voltage Ed across the smoothing capacitor Cs is based on perfect direct current. Actually, the voltage Ed across the smoothing capacitor Cs, has a ripple voltage Vpp, so that the voltage waveforms across the voltage-dividing capacitors C1 and C2 and the current waveform of the input current Ii become waveforms such as those shown in FIG. 25. Therefore, the voltage VR that is generated in the load circuit R has an uneven voltage waveform with both a maximum peak Vmax and a minimum peak Vmin, such as that shown in FIG. 26. Note that in FIG. 25, the voltages V.sub.C1 and V.sub.C2 are superposed and shown with alternating current zero as a boundary.
Notice that, as described above, in the neutral point inverter, by performing DCM (discontinuous mode) operation in which the input current Ii becomes a discontinuous current, there is no need to add an exclusive control circuit, which is required for preventing voltage rise during light load by other methods that perform CM (reactor current mode) or CRM (boundary mode) operation (e.g., an active filter circuit, etc.), unlike the neutral point inverter. Therefore, the neutral point inverter has the advantage that its circuitry becomes structurally simple.
For instance, the method that performs CM operation requires a detection-control circuit for detecting and controlling a position in which current flowing in an inductor does not go to zero. The method that performs CRM operation requires a detection-control circuit for detecting a position in which current through an inductor goes to zero and also detecting an output voltage to perform feedback control. On the other hand, the neutral point inverter does not require these exclusive circuits, and there is no possibility that voltage will rise so significantly even during light load. The voltage rise during light load has an influence upon the withstand voltage of used components, particularly an electrolytic capacitor for the smoothing capacitor CS, and switching elements such as field-effect transistors (FETs). For this reason, particularly in the case where a fluorescent lamp is used in the load, this state will occur each time the lighting of the fluorescent lamp is started and therefore what mode the input current Ii goes to is important.
Incidentally, to maintain stable lighting of a fluorescent lamp, the secondary voltage of a ballast is designed which is able to light the fluorescent lamp even under changes in external factors, such as fluorescent lamp characteristic dispersion, surrounding temperature and humidity change, and power-supply voltage fluctuation. On the other hand, in an independent ballast, for the wiring of a specific site of 300 V or less the code wiring is possible, whereas the secondary voltage that is high causes a disadvantage of wiring cost, and problems, such as an increase in the size of the ballast, safety, or a cold start countermeasure of a fluorescent lamp, will arise. Furthermore, in the case of a ballast sharing a plurality of fluorescent lamp types, such as slim-line fluorescent lamps, particularly a high-frequency secondary voltage design of low voltage which is able to reliably light lamps is required. For ordinary fluorescent lamps themselves, many narrow or long fluorescent lamps have also been used in recent years. This necessitates a high fluorescent lamp voltage. For example, in slim-line fluorescent lamps, the fluorescent lamp voltage is about 150 V, even for an intermediate size lamp (about 1 m), and in a normal state, an input voltage with an effective value of 200 V is required. Recently, with the spread of high-speed semiconductor sensors such as video displays, there is also a strong demand for a light source with less blinking. From these facts, a large difference between the maximum peak Vmax and minimum peak Vmin of the voltage supplied across a fluorescent lamp is disadvantageous. In consideration of efficiency, it is desirable that a fluorescent lamp be lit with even high-frequency voltage.
On the other hand, if the input voltage Vi does not require a 100-V system but a 200-V system, this means that an ordinary commercial power supply (100 V) cannot be used as it is. For this reason, there is a need to step up voltage with a transformer, etc., and handling becomes difficult. Also, the problem of safety will arise.
In addition, in the above-mentioned neutral point inverter, by inserting the low-pass filter LPF in the input thereof, an input current waveform is made approximately a current waveform which is nearly a sine wave, whereby harmonic current is prevented from flowing in a commercial power supply. However, there is a problem that the allowable range of the capacitance of the smoothing capacitor Cs is narrow and therefore there is no degree of freedom in selection.
The present invention has been made in view of the above-mentioned circumstances. Accordingly, it is an object of the present invention to provide a neutral point inverter type ballast or a neutral point inverter which is capable of obtaining a higher output and even (i.e., stable) output voltage and thereby rendering a reduction in an input voltage possible, while holding the characteristics of the neutral point inverter.